The present invention relates to a switching power-supply circuit having a power-factor improvement circuit.
The applicant for a patent of the present invention earlier proposed a variety of switching power-supply circuits each having a resonance-type converter on the primary side. In addition, there have been proposed a variety of switching power-supply circuits each having a power-factor improvement circuit for improving a power factor for the resonance-type converter.
FIG. 9 is a circuit diagram showing a typical switching power-supply circuit with a configuration based on an invention proposed earlier by the applicant for a patent of the present invention. To put it in detail, the configuration of this switching power-supply circuit includes a power-factor improvement circuit for improving the power factor of a switching converter of a current-resonance type based on a self-excitation technique.
The switching power-supply circuit shown in the figure includes a bridge rectifier circuit Di for full-wave rectification of the commercial AC power supply AC. A rectified output obtained as a result of the full-wave rectification by the bridge rectifier circuit Di is electrically charged into a smoothing capacitor Ci by way of a power-factor improvement circuit 20. As a result, a rectified and smoothed voltage Ei corresponding to a 1-time level of the AC input voltage VAC appears between the terminals of the smoothing capacitor Ci.
In addition, a rush-current limitation resistor Ri is inserted into a circuit comprising the bridge rectifier circuit Di and the smoothing capacitor Ci on a rectified-current path thereof. To put it in detail, the rush-current limitation resistor Ri limits a rush current flowing to the smoothing capacitor Ci when the power supply is turned on.
The power-factor improvement circuit 20 shown in the figure includes a filter choke coil LN and a high-speed recovery diode D1, which are connected to each other in series between a positive-electrode output terminal of the bridge rectifier circuit Di and the positive-electrode terminal of the smoothing capacitor Ci. One end of a choke coil LS is connected to the cathode of the high-speed recovery diode D1.
One terminal of a filter capacitor CN is connected to a connection between the anode of the high-speed recovery diode D1 and the positive-electrode terminal of the smoothing capacitor Ci. The other terminal of the filter capacitor CN is connected to the other end of the choke coil LS. The filter capacitor CN functions as a normal-mode low-pass filter in conjunction with the filter choke coil LN.
The connection point between the high-speed recovery diode D1 and the choke coil LS in the power-factor improvement circuit 20 is connected to a terminal of the primary side of a transformer PIT (Power Isolation Transformer) to be described later by a capacitor C1, which forms a series-resonance circuit in conjunction with an inductor L1 of a winding N1 on the primary side. With such a connection, a switching output generated by switching devices to be described later is fed back to the series-resonance circuit.
A power-factor improvement operation of the power-factor improvement circuit 20 will be described later.
The switching power-supply circuit also includes a converter of a current-resonance type adopting a self-excitation technique. This self-excitation current-resonance converter uses a rectified and smoothed voltage Ei appearing between the terminals of the smoothing capacitor Ci as an operation power supply.
As shown in the figure, the converter employs 2 switching devices Q1 and Q2 wired to each other in half-bridge connection between the positive-electrode terminal of the smoothing capacitor Ci and the ground to which the negative-electrode terminal of the smoothing capacitor Ci is connected. The switching devices Q1 and Q2 are each a bipolar transistor.
A start resistor RS1 is connected between the collector and the base of the switching device Q1. By the same token, a start resistor RS2 is connected between the collector and the base of the switching device Q2. A resistor RB1 connected to the base of the switching device Q1 through a resonance capacitor CB1 sets a base current (also referred to as a drive current) of the switching device Q1. Similarly, a resistor RB2 connected to the base of the switching device Q2 through a resonance capacitor CB2 sets a base current (also referred to as a drive current) of the switching device Q2. A clamp diode DD1 is connected between the emitter and the base of the switching device Q1. Likewise, a clamp diode DD2 is connected between the emitter and the base of the switching device Q2. The clamp diode DD1 forms a current path of a clamp current flowing through the base and the emitter of the switching device Q1 when the switching device Q1 is put in an off state. By the same token, the clamp diode DD2 forms a current path of a clamp current flowing through the base and the emitter of the switching device Q2 when the switching device Q2 is put in an off state.
The resonance capacitor CB1 forms a seriesresonance circuit for self-excitation oscillation in conjunction with a driving winding NB1 employed in a drive transformer PRT (power regulating transformer) to be described next, and sets the switching frequency of the switching device Q1. Likewise, the resonance capacitor CB2 forms a series-resonance circuit for self-excitation oscillation in conjunction with a driving winding NB2 employed in the drive transformer PRT, and sets the switching frequency of the switching device Q2. It should be noted that the series-resonance circuit is also referred to as a self-excitation oscillation driving circuit.
The drive transformer PRT drives the switching devices Q1 and Q2 as well as executes constant-voltage control by controlling variations in switching frequency. In the switching power-supply circuit shown in the figure, the driving windings NB1 and NB2, a resonance-current detection winding ND and a control winding NC oriented in a direction perpendicular to the driving windings NB1 and NB2 and the resonance-current detection winding ND form an orthogonal saturatable reactor.
One end of the driving winding NB1 employed in the drive transformer PRT is connected to the base of the switching device Q1 by a series connection of the resistor RB1 and the resonance capacitor CB1 whereas the other end of the driving winding NB1 is connected to the emitter of the switching device Q1. By the same token, one end of the driving winding NB2 employed in the drive transformer PRT is connected to the base of the switching device Q2 by a series connection of the resistor RB2 and the resonance capacitor CB2 whereas the other end of the driving winding NB2 is connected to the emitter of the switching device Q2. The driving windings NB1 and NB2 are wound in such directions that a voltage generated by the former has a polarity opposite to a voltage generated by the latter.
An insulating converter transformer PIT (Power Isolation Transformer) delivers outputs of the switching devices Q1 and Q2 on the secondary side. By connecting one end of the primary winding N1 of the insulating converter transformer PIT to a connection point (or a switching-output point) between the emitter of the switching device Q1 and the collector of the switching device Q2 through the resonance-current detection winding ND, a switching output is obtained.
As described above, the other end of the primary winding N1 is connected by the series-resonance capacitor C1 to a connection point between the cathode of the high-speed recovery diode D1 and the choke coil LS in the power-factor improvement circuit 20.
That is to say, the series-resonance capacitor C1 is connected in series to the primary winding N1. The capacitance of the series-resonance capacitor C1 and the leakage inductance of the insulating converter transformer PIT including the inductance L1 of the primary winding N1 form a primary-side series-resonance circuit for making the operation of the switching converter an operation of a current-resonance type. That is why the primary winding N1 is also referred to as a series-resonance winding.
On the secondary side of the insulating converter transformer PIT, a center tap is provided at the center of a secondary winding N2. The anodes of rectifier diodes D01 and D03 are connected to an upper-end tap and an upper middle tap of the secondary winding N2 respectively. By the same token, the anodes of rectifier diodes D02 and D04 are connected to a lower-end tap and a lower middle tap of the secondary winding N2 respectively. A smoothing capacitor C01 is connected between the ground and the cathodes of the rectifier diodes D01 and D02 to form a first full-wave rectification circuit. Likewise, a smoothing capacitor C02 is connected between the ground and the cathodes of the rectifier diodes D03 and D04 to form a second full-wave rectification circuit. The first full-wave rectification circuit comprising the smoothing capacitor C01 and the rectifier diodes D01 and D02 generates a direct-current output voltage E01. Similarly, the second full-wave rectification circuit comprising the smoothing capacitor C02 and the rectifier diodes D03 and D04 generates a direct-current output voltage E02.
It should be noted that the direct-current output voltage E01 and the direct-current output voltage E02 are supplied separately to a control circuit 1. The control circuit 1 uses the direct-current output voltage E01 as a detection voltage and the direct-current output voltage E02 as an operation power supply.
The control circuit 1 executes constant-voltage control to be described later. To put it concretely, the control circuit 1 supplies a DC current to the control winding NC of the drive transformer PRT as a control current. Typically, the magnitude of the control current is adjusted in accordance with variations in direct-current output voltage E01 on the secondary side.
The switching power-supply circuit with a configuration described above carries out a switching operation as follows. First of all, when the commercial AC power supply is turned on, activation currents are supplied to the bases of the switching devices Q1 and Q2 by way of the start resistors RS1 and RS2. Assuming that the switching device Q1 is turned on earlier, control is executed to turn off the switching device Q2. As an output of the switching device Q1, a resonance current flows to the resonance-current detection winding ND, the primary winding N1 and the series-resonance capacitor C1. Control is executed to turn off the switching device Q1 but turn on the switching device Q2 as the magnitude of the resonance current approaches 0. This time, as an output of the switching device Q2, a resonance current flows in a direction opposite to the resonance current generated as the output of the switching device Q1. Thereafter, a self-excitation switching operation wherein the switching device Q1 and the switching device Q2 are turned on alternately is started.
As described above, the switching device Q1 and the switching device Q2 are alternately and repeatedly turned on and off with the voltage between the terminals of the smoothing capacitor Ci used as an operation power supply. As a result, a drive current having a waveform close to a resonance-current waveform is supplied to the primary winding N1 of the insulating converter transformer PIT and an AC output is obtained at the secondary winding N2 thereof.
The constant-voltage control cited earlier is executed by the drive transformer PRT as follows.
Assuming that the secondary-side output voltage E01 increases due to a change in AC input voltage and/or a change in load, the control current flowing through the control winding NC is also controlled to rise in accordance with the increase in the secondary-side output voltage E01.
Due to an effect of a magnetic flux generated by this control current in the drive transformer PRT, the drive transformer PRT approaches a saturated state, exhibiting an effect of decreasing the inductances of the driving windings NB1 and NB2. Thus, the condition of the self-excitation resonance circuit changes, increasing the switching frequency.
In this switching power-supply circuit, upper-side control is executed. That is to say, the switching frequency is set at a value in a frequency region higher than the resonance frequency of the series-resonance circuit, which comprises the series-resonance capacitor C1 and the inductor L1 of the primary winding N1. As the switching frequency is controlled to rise as described above, the switching frequency departs from the resonance frequency of the series-resonance circuit. As a result, the resonance impedance of the series-resonance circuit for the switching output increases.
When the resonance impedance increases as described above, a drive current supplied to the primary winding N1 of the series-resonance circuit on the primary side is limited. As a result, the output voltage appearing on the secondary side is also limited by constant-voltage control.
The constant-voltage control based on the technique described above is referred to hereafter as a switching-frequency control method.
The power-factor improvement circuit 20 carries out a power-factor improvement operation as follows.
In the configuration of the power-factor improvement circuit 20 shown in the figure, the switching output supplied to the series-resonance circuit comprising the inductor L1 of the primary winding N1 and the series-resonance capacitor C1 is fed back to the rectified-current path by way of an inductive reactance (or magnetic coupling) of the choke coil LS itself.
The switching output fed back as described above causes an alternating voltage having a switching period to be superposed on the rectified-current path. The superposition of the alternating voltage having the switching period in turn causes a rectified current to flow through the high-speed recovery diode D1 intermittently at the switching period. The intermittent flow of the rectified current causes the inductances of the filter choke coil LN and the choke coil LS to appear higher. Thus, also during a period in which the level of the rectified output voltage is lower than the voltage appearing between the terminals of the smoothing capacitor Ci, a charging current flows to the smoothing capacitor Ci.
As a result, the average waveform of the AC input current approaches the waveform of the AC input voltage and the conduction angle of the AC input current increases to improve the power factor.
FIG. 10 is a circuit diagram showing another typical configuration of the switching power supply circuit with a configuration based on the present invention proposed earlier by the applicant for a patent of the present invention. This switching power-supply circuit also includes a current-resonance converter in which 2 switching devices are wired to form a half-bridge junction. A separate-excitation technique is adopted as a driving method. The configuration of this switching power-supply circuit also includes a power-factor improvement circuit for improving the power factor.
It should be noted that components identical with those employed in the switching power-supply circuit shown in FIG. 9 are denoted by the same reference numerals as the latter and their explanation is not repeated.
As shown in the figure, the current-resonance converter on the primary side employs 2 switching devices Q11 and Q12, which are each implemented by typically a MOS-FET.
The drain of the switching device Q11 is connected to the line of a rectified and smoothed voltage E1. The source of the switching device Q11 is connected to the drain of the switching device Q12. The source of the switching device Q12 is connected to the ground on the primary side. With such connections, a half-bridge junction associated with the separate-excitation technique is resulted in.
The switching devices Q11 and Q12 are driven by an oscillation and drive circuit 2 to turn on and off alternately and repeatedly in switching operations to output the rectified and smoothed voltage Ei intermittently.
A clamp diode DD1 is connected between the drain and the source of the switching device Q11 in a direction shown in the figure. By the same token, a clamp diode DD2 is connected between the drain and the source of the switching device Q12 in a direction shown in the figure.
By connecting one end of the primary winding N1 of the insulating converter transformer PIT to the connection point (also referred to as a switching output point) between the source of the switching device Q11 and the drain of the switching device Q12, the switching output can be supplied to the primary winding N1. The other end of the primary winding N1 is connected to the connection point between the filter choke coil LN and the anode of the high-speed recovery diode D1 in the power-factor improvement circuit 21 to be described later.
Also in the case of the switching power-supply circuit shown in FIG. 10, the series-resonance capacitor C1 is connected in series to the primary winding N1. The capacitance of the series-resonance capacitor C1 and the leakage inductance of the insulating converter transformer PIT including the inductance of the primary winding N1 form a primary-side series-resonance circuit for making the operation of the switching power-supply circuit an operation of a current-resonance type.
The control circuit 1 of this configuration outputs a control signal with a level representing typically a variation in direct-current output voltage E01. In the oscillation and drive circuit 2, the frequencies of switching driving signals supplied by the oscillation and drive circuit 2 to the gates of the switching devices Q11 and Q12 are varied in accordance with the control signal received from the control circuit 1 in order to change the switching frequency.
Also in the switching power-supply circuit shown in FIG. 10, the switching frequency is set at a value in an area higher than the series-resonance frequency as is the case with the switching power-supply circuit shown in FIG. 9. When the direct-current output voltage E01 rises, for example, the oscillation and drive circuit 2 is controlled by the control circuit 1 so that the switching frequency also increases in accordance with the level of the direct-current output voltage E01, therby to execute the constant voltage control.
A start circuit 3 detects a voltage or a current on the rectified and smoothed line and activates the oscillation and drive circuit 2 right after the power supply is turned on. A low-level direct-current voltage obtained by rectifying a winding additionally provided in the insulating converter transformer PIT is supplied to the start circuit 3 as an operation power supply.
The power-factor improvement circuit 21 shown in the figure includes a filter choke coil LN and a high-speed recovery diode D1, which are connected to each other in series between a positive-electrode output terminal of the bridge rectifier circuit Di and the positive-electrode terminal of the smoothing capacitor Ci. A filter capacitor CN is connected in parallel to the series connection circuit comprising the filter choke coil LN and the high-speed recovery diode D1. Also in this connection, the filter capacitor CN functions as a normal-mode low-pass filter in conjunction with the filter choke coil LN.
A resonance capacitor C3 is connected in parallel to the high-speed recovery diode D1. Typically, the resonance capacitor C3 forms a parallel-resonance circuit in conjunction with a component such as the filter choke coil LN. The parallel-resonance circuit is set to have a resonance frequency about equal to the resonance frequency of a series-resonance circuit to be described later. In this way, there is exhibited an effect of suppression of an increase in rectified and smoothed voltage Ei caused by a reduced load. No more detailed description is given.
As described earlier, the connection point between the filter choke coil LN and the anode of the high-speed recovery diode D1 in the power-factor improvement circuit 21 is connected to the aforementioned series-resonance circuit comprising an inductor L1 of the primary winding N1 and the series-resonance capacitor C1.
In the connection described above, a switching output obtained at the primary winding N1 is fed back to the rectified-current path by way of electrostatic-capacitance coupling of the series-resonance capacitor C1. To put it in detail, the switching output is fed back so that a resonance current obtained at the primary winding N1 flows to the connection point between the filter choke coil LN and the anode of the high-speed recovery diode D1, applying the switching output to the connection point.
The switching output fed back as described above causes an alternating voltage having a switching period to be superposed on the rectified-current path. The superposition of the alternating voltage having the switching period in turn causes a rectified current to flow through the high-speed recovery diode D1 intermittently at the switching period. The intermittent flow of the rectified current causes the inductances of the filter choke coil LN to appear higher.
In addition, since a current with the switching period flows through the resonance capacitor C3, a voltage appears between the terminals of the resonance capacitor C3. The level of the rectified and smoothed voltage Ei decreases by an amount equal to the voltage appearing between the terminals of the resonance capacitor C3. Thus, even during a period in which the rectified output voltage level is lower than the voltage appearing between the terminals of the smoothing capacitor Ci, a charging current flows into the smoothing capacitor Ci.
As a result, the average waveform of the AC input current approaches the waveform of the AC input voltage and the conduction angle of the AC input current increases to improve the power factor as is the case with the switching power-supply circuit shown in FIG. 9.
By providing the switching power-supply circuits shown in FIGS. 9 and 10 with the power-factor improvement circuits 20 and 21 respectively as described above, the power factor can be improved. Since the power-factor improvement circuits 20 and 21 shown in these figures each employ a small number of components, the power-factor improvement circuits 20 and 21 each offer merits that the power factor can be improved with a high degree of efficiency, at a small amount of noise, by using a circuit with a small size and a small weight and at a low cost.
FIG. 11 is a diagram showing a relation between the load power Po and the power factor PF for the switching power-supply circuits shown in FIGS. 9 and 10. It should be noted that an AC input voltage VAC of 100 V is set as a condition.
The relation shown in the figure represents a characteristic of the power factor PF decreasing with reductions in load power Po as is obvious from the figure.
FIG. 12 is a diagram showing relations between the AC input voltage VAC and the power factor PF. These relations represent characteristics with a maximum load power Pomax of 120 W and a minimum load power Pomin of 40 W set as conditions.
As is obvious from the figure, the power factor PF decreases proportionally with increases in AC input voltage VAC.
In addition, the power factor PF for the minimum load power Pomin of 40 W is lower than the power factor PF for the maximum load power Pomax of 120 W. This relation agrees with the characteristic of FIG. 11, which shows a lower power factor PF for a smaller load power Po.
FIGS. 13A to 13D are diagrams showing operation waveforms for the characteristics shown in FIG. 12.
To be more specific, FIG. 13A is a diagram showing the waveform of the AC input voltage VAC for an AC input voltage VAC of 100 V and at a maximum load power Pomax of 120 W. FIG. 13B is a diagram showing the waveform of the AC input current IAC for an AC input voltage VAC of 100 V and at a maximum load power Pomax of 120 W. FIG. 13C is a diagram showing the waveform of the AC input voltage VAC for an AC input voltage VAC of 100 V and at a minimum load power Pomin of 40 W. FIG. 13D is a diagram showing the waveform of the AC input current IAC for an AC input voltage VAC of 100 V and at a minimum load power Pomin of 40 W.
Assuming that the half period of the AC input voltage VAC is 10 ms, at the maximum load power Pomax of 120 W, the conduction period .tau. of the AC input current IAC is actually about 5 ms and the power factor PF is thus 0.85. At the minimum load power Pomin of 40 W, on the other hand, the conduction period .tau. of the AC input current IAC decreases to about 2.5 ms and the power factor PF is also reduced to about 0.65. The value of the power factor PF for the minimum load power Pomin of 40 W may not be a satisfactory value of a power factor PF required by some applications.
A decrease in power factor caused by a change in AC input voltage and/or a change in load power, conversely speaking, means a limitation on the AC input voltage condition and/or the load power condition for the switching power-supply circuit. That is to say, there is raised a problem of limited kinds of equipment that can employ the switching power-supply circuit.
To put it concretely, while a switching power-supply circuit can be employed in a television receiver with specified AC input voltage and/or the load power conditions, the same switching power-supply circuit may not be usable in office or information equipment.
In addition, in the configurations of FIGS. 9 and 10 for improving the power factor, the series-resonance circuit on the primary side is connected to the rectified-current path of the commercial AC power supply. As a result, ripples with the commercial AC power-supply frequency of 50 or 60 Hz are superposed on the series-resonance circuit as is generally known. The superposition level of the ripple component becomes higher with an increase in load power.
Assuming that the configuration includes components that are selected to maintain the power factor PF at about 0.8 as measured under predetermined conditions for an application, as is generally known, the voltage level of ripples appearing in the direct-current output voltage on the second side at a maximum load power increases by about 3 to 4 times in comparison with a case including no power-factor improvement circuit.
In order to suppress the increase in ripple component level described above, in the actual implementation of the switching power-supply circuits shown in FIGS. 9 and 10, the gain of the control circuit 1 and/or the capacitance of the smoothing capacitor Ci on the primary side are increased. In this case, however, there will be problems of an increased component cost and a switching operation prone to oscillation.